Electric motor

ABSTRACT

A variable motor ( 1 ) having reduced audible noise. This advantage is achieved by reducing or eliminating ripple torque by making the individual phase torques follow a certain profile, so that when the torques are summed, they are substantially constant. Therefore, the motor ( 1 ) back EMF is used as a reference waveform to control the winding currents to achieve reduced ripple torque. The back EMF reference is amplified and provided to the motor windings, and may be derived in a number of ways including use of sense windings, electronic derivation, and approximation by sensing the voltage applied to the windings.

FIELD OF THE INVENTION

This invention relates to electric motors in particular, but notexclusively to, variable speed brushless electric motors.

BACKGROUND ART

Variable speed electric motors are known prior art. In many motorapplications audible noise is undesirable, and can become particularlynoticeable at low motor speeds and when the electrical supply to suchmotors is varied during transitions from the original speed to thevaried speed. Audible noise is also associated with energy loss. It isalso known that speed variations of prior art motors are difficult tocontrol because these motors are generally designed to operate at afixed supply frequency. Speed control difficulties are most pronouncedin prior art fractional and subfractional horsepower motors with designswhich have been optimised or minimised for low manufacturing costs. Itis however possible to improve the speed variation and control of thesemotors by electronic means. In addition mechanical means such as but notlimited to gearboxes, belts, variators and electromagnetic couplings mayalso be incorporated to improve speed control. As a consequence, theseimprovements increase the cost and complexity of the complete motorassembly. Prior an motors have also been found to have energyefficiencies (defined as the ratio of mechanical output power toelectrical input power) of 20% or less when operating at fixed speeds.The addition of speed control means reduces the energy efficiency stillfurther.

OBJECT OF THE INVENTION

It is one object of the present invention to provide a variable speedelectric motor which is more efficient or audibly quieter at least whenvarying speeds, or to at least provide the public with a useful choice.

SUMMARY OF THE INVENTION

According to one aspect the invention provides an electric motorincluding means for determining a first signal associated with the rotor(when the motor is operational) and wherein said first signal is appliedas an input signal to a power amplifier, which generates an output whichis applied to windings of the motor to drive the motor, and whereby inuse, audible noise associated with the motor is thus significantlyreduced.

The invention may make use of a sense winding as described in ourearlier New Zealand Patent application 501086 (the contents of which areincorporated herein by way of reference) to determine the first signal,or it may simulate this signal electronically without the need for asense winding. As a further alternative the invention may derive anapproximation of the first signal by sensing the voltage or currentapplied to the windings. In either case the first signal is applied to apower amplifier to create an output which is applied to the motor todrive the motor.

According to a second aspect the invention provides an electric motorincluding electronic back EMF sensing means, the electronic back EMFsensing means including voltage sensing means and current sensing meansand rate-of-change of current sensing means adapted to sense a back EMFgenerated by a rotor of the motor wherein the sensed electronic back EMFis applied as an input signal to a power amplifier, which generates anoutput which is applied to windings of the motor as a drive voltage orcurrent, and whereby in use, audible noise associated with the motor isthus significantly reduced.

According to a third aspect the invention provides an electric motorincluding sensing means to sense an approximation of the back EMFproduced by rotation of the rotor by sensing the voltage or currentapplied to a motor winding, the sensing means providing a first signalwhich is applied as an input signal to a power amplifier, whichgenerates an output which is applied to windings of the motor as a drivevoltage or current, and whereby in use, audible noise associated withthe motor is thus significantly reduced.

According to a fourth aspect the invention provides an electric motorincluding back EMF prediction means, the back EMF prediction meansincluding a memory from which data relating to the motor are stored, theback EMF prediction means providing a first signal indicative of themotor back EMF, the first signal being applied as an input signal to apower amplifier, which generates an output which is applied to windingsof the motor as a drive voltage or current, and whereby in use, audiblenoise associated with the motor is thus significantly reduced.

Preferably the electric motor is a radial flux permanent magnet motor.In the alternative the electric motor can be an axial flux motor orother type of direct current motor.

Preferably the motor is a brushless motor.

Preferably the rotor is an external rotor rotatable about a centralshaft.

Preferably the rotor houses a plurality of spaced permanent magnetsarranged in a radial pattern on an internal surface of the rotor.

Preferably the electronic back EMF sensing means includes currentsensing means which detect the steady state value of the motor windingcurrent in each motor phase.

Preferably the electronic back EMF sensing means also includesrate-of-change of current sensing means which detect the rate-of-changeof motor winding current in each phase.

Preferably the electronic back EMF sensing means also includesvoltage-sensing means which sense the voltage applied to the motorwindings.

Preferably the electronic back EMF sensing means also includes a summingamplifier which allows the addition of signals from the current sensingmeans and the rate-of-change of current sensing means and the voltagesensing means, so that the output of the summing amplifier is analogousto the voltage field generated by the rotor of the motor.

Preferably the electronic back EMF sensing means also includes filteringcircuits that can introduce a phase shift between the summing amplifierinput voltages and the summing amplifier output voltage which is thenapplied to the power amplifier input.

Preferably the power amplifier means comprises power conversion means toadjust an available supply voltage to voltage levels suitable fordriving the motor at the required speed controlled either directly orindirectly by the electronic back EMF sensing means.

In another aspect the invention resides in a method of reducing audiblenoise associated with an electric motor including the steps of:

1. Sensing a first signal associated with the voltage generated by amotor's rotor from rotation of the rotor in use.

2. Amplifying and applying directly the first signal to windings of themotor as a drive voltage or current.

3. Controlling the drive voltage or current amplitude to control themotor whereby in operation audible noise is significantly reduced.

Preferably the electric motor is a radial flux permanent magnet motor.In the alternative the electric motor can be an axial flux motor orother type of direct current motor.

Preferably the motor is a brushless motor.

Preferably the rotor is an external rotor rotatable about a centralshaft.

Preferably the rotor houses a plurality of spaced permanent magnetsarranged in a radial pattern on an internal surface of the rotor.

Preferably the search coil means comprises a coil or coils woundcoincidentally with one or more of the drive windings of the motor.

Preferably the electronic field sensing means includes a power amplifierthat amplifies the search coil voltage. Preferably the electronic fieldsensing means also includes filtering circuits that can introduce aphase shift between the search coil voltage and the power amplifieroutput voltage which is then applied to the motor windings.

Preferably the power amplifier means comprises power conversion means toadjust an available supply voltage to voltage levels suitable fordriving the motor at the required speed controlled either directly orindirectly by the electronic field sensing means.

In another aspect the invention resides in a method of reducing audiblenoise associated with an electric motor including the steps of:

1. Sensing, approximating or predicting a voltage or current generatedby a motor's rotor.

2. amplifying and applying directly the generated voltage or current towindings of the motor as a drive voltage or current,

3. controlling the drive voltage or current amplitude to control themotor whereby in operation audible noise is significantly reduced.

BRIEF DESCRIPTION OF THE DRAWINGS

In order that the present invention can be more readily understood andput into practical effect, reference is made to the accompanyingdrawings of the preferred embodiments wherein:

FIG. 1: is a plan view of a rotor and stator of a preferred motoraccording to Example 1;

FIG. 2: is a plan view of a rotor and stator of a preferred motoraccording to Example 2, and

FIG. 3: shows a schematic diagram representing a preferred circuitarrangement of the invention.

FIG. 4: is a plan view of a rotor and stator of a preferred motoraccording to Example 4;

FIG. 5: is a plan view of a rotor and stator of a preferred motoraccording to Example 5,

FIG. 6: shows a schematic diagram representing a preferred circuitarrangement of the invention, and

FIG. 7: shows a schematic diagram representing a further preferredcircuit arrangement of the invention.

DETAILED DESCRIPTION OF EXAMPLES OF THE INVENTION

The invention achieves reduced acoustic noise (and thus also preferablyreduces energy losses) by reducing or substantially eliminating rippletorque. This can be achieved by making the individual phase torquesfollow a certain profile, so that when the torques are summed, they aresubstantially constant. For example, in a 2 phase motor with 90 degreedisplacement between windings, zero ripple torque is generated when theindividual phase torques follow a sin²(θe) profile, where θe is theangular position of the rotor magnetic field.

Where the motor is designed so that the rotor flux linkage with thewindings is a sinusoidal function of θe, and the winding currents arealso sinusoidal function of θe, then the torque contribution for asingle phase is proportional to sin²(θe). Thus, again in the example ofa two phase motor, when there is sinusoidal winding flux linkage and 90degree phase displacement between windings there is zero ripple torqueif the instantaneous winding currents are in proportion to theinstantaneous flux linkage in each phase.

The motor back EMF is proportional to the flux linkage. Therefore, theinvention embodies use of the motor back EMF as a reference waveform tocontrol the winding currents.

EXAMPLE 1

FIG. 1 of Example 1 depicts a plan view of a preferred external rotor 4pole 2 phase brushless dc motor (1). Alternatively the number of motorpoles may be any even number in the range 2-40. Alternatively the numberof motor phases may be any number in the range 1-12. An 8 slot statorassembly (10) is located concentric to a rotor assembly (12) The rotorassembly has 4 magnetic poles (11, 13, 15, 17) arranged radially on theinternal surface of the rotor.

The drawing shows the location of an A phase search coil (14) and the Aphase main winding (16). The search coil consists of 20 turns spanning asingle stator pole (not shown). The main winding consists of 40 turnsper pole spanning 4 stator poles (not shown). The full number of turnsare not shown on the drawing for reasons of clarity, however theconductor paths are clearly shown. The winding start and end points arelabelled as follows:

(i) A-phase search coil start (18);

(ii) A-phase search coil end (22);

(iii) A-phase main winding start (20);

(iv) A-phase main winding end (24).

EXAMPLE 2

In this example, like parts are given the same numbering as these ofFIG. 1

FIG. 2 of Example 2 depicts a plan view of a preferred external rotor 4pole 2 phase brushless dc motor (2). The stator (10) and rotor (12)assembly showing in addition to FIG. 1, the location of the B phasesearch coil (26) and the B-phase main winding (28). The B phase windingstart and end points are labelled as follows:

(i) B-phase search coil start (32);

(ii) B-phase search coil end (36);

(iii) B-phase main winding start (30);

(iv) B-phase main winding end (34).

FIG. 3:

Shows a schematic diagram (3) of the circuit of the preferred embodimentof the electronic field sensing means and power amplifier means. Vsearch(52) represents the electro motive force (emf) induced in the searchcoils (not shown) as the rotor magnetic field moves over them. Themagnitude of Vsearch is proportional to the rotor revolution per minute(rpm) and has a voltage coefficient, Kvs, of 1V/1000 rpm. Vs (54)represents the voltage measured between (18) and (22). The electronicfield sensing circuit is designed to present negligible electrical loadto the search coil, so that Vs and Vsearch can be regarded asequivalent.

Vbemf (58) represents the emf induced in the main windings (not shown)as the rotor magnetic field moves over them. The magnitude of Vbemf isproportional to the rotor rpm and has a voltage coefficient, Kvm, of8V/1000 rpm. Vapp (56) represents the voltage measured between (20) and(24). Iw (60) represents the current flowing in the main winding (notshown). If no current is flowing in the motor winding i.e. Iw=0, thenVapp=Vbemf. For any non-zero value of Iw, Vapp and Vbemf will differ dueto internal voltage drop within the main winding.

Vs is passed through a filter stage (40) to remove spurious noise andintroduce a phase shift if required. The filter output signal is passedthrough a voltage amplifier A1 (42) and then into a power amplifier A2(48). The power amplifier draws electrical power from an availablesupply (50) and applies the amplified search coil voltage, Vapp, to themain winding. The overall voltage gain, Av, of amplifiers A1 (42) and A2(48) operating in series can be defined as the ratio of Vapp divided byVs.

In order to generate motoring torque in the motor, the magnitude of Vappmust be greater than vbemf In order to achieve this Av must be greaterthan Kvm divided by Kvs i.e. for the windings shown Av must be greaterthan 8.

Alternatively, if Av is set so that Vapp is lower than Vbemf, then Iwwill flow in the reverse direction and the motor will generate a brakingtorque. Thus, the magnitude and direction of motor output torquegenerated at a given rpm is determined by the value of Av.

In the embodiment shown, all of the system voltage gain is implementedin amplifier A1. Amplifier A2 is a unity gain voltage follower, whichprovides current amplification to supply the stator winding.Alternatively, amplifiers A1 and A2 may be implemented in a singleintegrated amplifying device, such as an amplifier integrated circuit.

In the embodiment shown, the voltage gain of A1 is set by the value ofvariable resistor VR1 (46), divided by resistor R1 (44). Therefore,adjusting the value of VR1 varies Av, which controls the magnitude anddirection of motor torque at a given rpm. Alternatively Av may be variedby use of an electronic gain control circuit (not shown). Alternatively,Av may be constant and Vapp controlled by means of variable attenuationof Vs. Alternatively, any circuit means which allows variation of Av asdefined above may be employed.

EXAMPLE 4

FIG. 4 of Example 4 depicts a plan view of a preferred external rotor4-pole 2-phase brushless dc motor (101).

Alternatively the number of motor poles may be any even number in therange 2-40. Alternatively the number of motor phases may be any numberin the range 1-12.

An 8-slot stator assembly (110) is located concentric to a rotorassembly (112). The rotor assembly has 4 magnetic poles (111, 113, 115,117) arranged radially on the internal surface of the rotor.

The drawing shows the location of the A phase main winding (116). Themain winding consists of 40 turns per pole spanning 4 stator poles (notshown). The full number of turns are not shown on the drawing forreasons of clarity, however the conductor paths are clearly shown. Thewinding start and end points are labelled as follows:

(i) A-phase main winding start (120);

(ii) A-phase main winding end (124).

EXAMPLE 5

In this example, like parts are given the same numbering as these ofFIG. 4.

FIG. 5 of Example 5 depicts a plan view of a preferred external rotor4-pole 2-phase brushless dc motor (102). The stator (110) and rotor(112) assembly showing in addition to FIG. 1, the location of theB-phase main winding (128). The B phase winding start and end points arelabelled as follows:

(i) B-phase main winding start (130);

(ii) B-phase main winding end (134).

EXAMPLE 6

FIG. 6 shows a schematic diagram of the circuit of the preferredembodiment of the electronic back EMF sensing means and power amplifiermeans.

Vbemf (158) represents the EMF induced in the main windings (not shown)as the rotor magnetic field moves over them. The magnitude of Vbemf isproportional to the rotor rpm and has a voltage coefficient, Kvm, of8V/1000 rpm. Rw (198) represents the resistance of the motor winding,measured between terminals (120) and (124). Lw (200) represents theinductance of the motor windings measured between terminals (120) and(124). Vapp (156) represents the voltage applied to the motor windingsmeasured between (120) and (202). Iw (160) represents the currentflowing in the main winding (not shown). If no current is flowing in themotor winding i.e. Iw=0, then Vapp=Vbemf. For any non-zero value of Iw,Vapp and Vbemf will differ due to internal voltage drop within the mainwinding.

Electric current flowing in the motor winding generates a voltage, Vs(154), across a current sensing resistor, Rs (164). Rs is chosen to besignificantly lower resistance than Rw (Rs<0.1Rw) so that negligibleerror is incurred in considering Vapp to be equivalent to the voltagemeasured between (120) and (124). Vs is connected to the input ofcurrent detection circuit comprising amplifier A1(166) and associatedcomponents R168 (168), R170 (170), R174 (174), C172 (172). The voltagegain, Av1, of amplifier A1 is set so that:

Rs multiplied by Av1=Rw,

so that A1 output voltage, VRw(204) is equal in magnitude to the voltagedrop across Rw due to Iw.

A1 is an inverting amplifier, so VRw=−(Rw×Iw).

Av1 is equal to the ratio of R170 divided by R168 and this ratio mustthus be equal to Rw divided by Rs.

Capacitor C172 is selected to reduce Av1 at frequencies above themaximum motor operating frequency.

R174 is chosen to be the standard resistor value closest to the value ofR168 and R170 in parallel combination, according to standard designpractise.

Vs is also connected to the input of the rate-of-change of currentdetection circuit comprising amplifier A2 (176) and associatedcomponents C178 (178), C184 (184), R180 (180), R182 (182), R186 (186).C178 and R182 are chosen so that A2 output voltage, VLw (206) is equalin magnitude to the rate-of-change of Iw through Rs multiplied by Lw. A2is an inverting amplifier, so VLw=−(Lw×d(Iw)/dt).

R180 and C184 are added to limit the response of the circuit to noise atfrequencies above the maximum motor operating frequency, according tostandard design practise.

A3 (188) forms a summing amplifier, where Vapp, VRw and VLw are added toform a composite voltage signal Vbemf* (162).

Resistors R190, R192, R194 and R196 allow the relative weighting ofVapp, VRw and VLw to be scaled if required.

The output of A3 is described by the equation; Vbemf*=−(Vapp−VRw−VLw).

By comparison it can be seen that this is analogous to the actual motorback emf voltage, which is described by the equationVbemf=(Vapp−(Iw×Rw)−(Lw×d(Iw)/dt)).

Vbemf* is applied to the input of power amplifier A4 (148).

The system voltage gain, Avs, can be defined as the ratio of Vappdivided by Vbemf. The voltage gain of A4 can be defined as the ratio ofVapp divided by Vbemf*. The power amplifier draws electrical power froman available supply (150) and applies an output voltage, Vapp,proportional to Vbemf*, to the main winding. In order to generatemotoring torque in the motor, the amplitude of Vapp must be greater thanVbemf. In order to achieve this Avs must be greater than unity, or whereVbemf* is scaled so as to be equal in magnitude to Vbemf, then Av4 mustbe greater than unity.

Alternatively, if Avs is set so that Vapp is lower than Vbemf, then Iwwill flow in the reverse direction and the motor will generate a brakingtorque. Thus, the magnitude and direction of motor output torquegenerated at a given rpm is determined by the value of Avs.

In the embodiment shown, the electronic back emf sensing circuit output,Vbemf*, is equal in magnitude to the motor back emf, Vbemf and all ofthe system voltage and current gain is implemented in power amplifierA4, so that Av4 is equivalent to Avs. Alternatively any combination ofvoltage and current amplification stages may be employed.

In the embodiment shown, Av4 is set by the value of variable resistorVR1 (146), divided by resistor R1 (144). Therefore, adjusting the valueof VR1 varies Avs, which controls the magnitude and direction of motortorque at a given rpm. Alternatively Avs may be varied by use of anelectronic gain control circuit (not shown). Alternatively, Avs may beconstant and Vapp controlled by means of variable attenuation of Vbemf*.Alternatively, any circuit means which allows variation of Avs asdefined above may be employed.

EXAMPLE 7

FIG. 7 shows a schematic diagram (1) of the circuit of the preferredembodiment of the electronic applied voltage sensing means and poweramplifier means.

Vbemf (58) represents the emf induced in the main windings (not shown)as the rotor magnetic field moves over them. The peak magnitude of Vbemfis proportional to the rotor rpm and has a voltage coefficient, Kvm, of8V/1000 rpm. Vbemf is an alternating current waveform, with frequency,fcom, proportional to rpm and number of motor magnetic poles asdescribed by equation; fcom (Hz)=rpm/60×(number of motor poles)/2.

Rw (98) represents the distributed resistance of the motor winding,measured between terminals (20) and (24). Lw (100) represents thedistributed inductance of the motor windings measured between terminals(20) and (24) Xw (not shown) represents the reactance of Lw at fcom, asdescribed by equation, Xw=2π×fcom×Lw. Zw (not shown) represents thecombined ac impedance of Rw and Xw at fcom, as described by equation;Zw=(Rw²+Xw²).

Vapp (56) represents the voltage measured between (20) and (24). Iw (60)represents the current flowing in the main winding (not shown). Vw (236)represents the voltage drop across Zw due to Iw. Vw is internal to themotor and cannot be directly measured by physical means. Vw can becalculated from equation; Vw=Zw×Iw. Vapp can also be calculated fromequation; Vapp=Vbemf+(Iw×Zw).

Vapp is filtered and attenuated by the Vapp sensing circuit comprisingcomponents R200 (200), VR202 (202), C204 (204). KVapp(206) representsthe voltage measured between (234) and (20), which is inverted inpolarity with respect to Vapp. R200 and VR202 form a potential dividerwhich attenuates Vapp by a fraction K, so that KVapp=−K×Vapp, where Kcan be altered by changing the setting of VR202. C204 provides filteringof the KVapp signal to remove spurious circuit noise and introduce aphase delay between KVapp and Vapp.

KVapp is connected to the input of the winding current control circuitcomprising amplifier A216 (216) and associated components R208 (208),R210 (210), 214 (214), C212 (212). A216 output is connected to the inputof a power amplifier circuit comprising T222 (222), T224 (226), R220(220), R230 (230), C224 (224), C232 (232). The power amplifier drawspower from an available supply (50) and applies power to the motorwinding.

Vs (228) represents voltage measured between (218) and (20). Vs is equalto the voltage drop across R230 due to current flowing in the motorwindings, i.e. Vs=Iw×R230. A216 is configured as an inverting amplifier,with voltage gain, Av equal to the ratio of R208 and R214; i.e.Av=−(R214/R208). Negative voltage feedback is provided by connectingfeedback network R214, C212 to Vs. C212 is selected to reduce Av atfrequencies above maximum fcom. R210 is chosen to be a standard resistorvalue close to the value R208 and R214 in parallel combination,according to standard design practise. C232 provides energy absorptionto limit the rate-of-rise of voltage Vapp.

As Vs is proportional to Iw, the winding current control circuit andpower amplifier circuit acting in series form a transconductanceamplifier where the output current, Iw, is proportional to the inputvoltage, KVapp. For a given setting of VR202 KVapp is proportional toVapp so the transconductance amplifier varies Iw in proportion to Vapp.The magnitude of Iw is controlled by altering the setting of VR202. Theratio of Iw to Vapp can be defined as the system transconductance Gs(not shown), described by equation Gs=Iw/Vapp.

The relationship between Vapp, Vbemf, Iw and Zw is described by thecharacteristic equation for a dc machine; Vapp=(Iw×Zw)+Vbemf

By substituting (Gs×Vapp) for Iw, and rearranging it can be shown that:

Vapp=Vbemf×(I−Gs×Zw)⁻¹

So, since Vapp is proportional to Vbemf, and Iw is proportional to Vapp,it follows that for the control system described above Iw isproportional to Vbemf for a given Gs.

In the embodiment shown the transconductance amplifier has fixed gainand Gs is varied via VR202. Alternatively Gs may be varied by alteringthe gain of the transconductance amplifier. Alternatively Gs may bevaried by the inclusion of additional attenuation networks (not shown).Alternatively Gs may be varied by the use of an electronic gain controlcircuit (not shown). Alternatively, any circuit means which allowsvariation of Gs as defined above may be employed.

In the embodiment shown, a transconductance amplifier is formed by awinding current control stage in series with a power amplifier stage.Alternatively, a transconductance amplifier comprising any combinationof signal and power amplification stages may be employed.

In the embodiment shown the circuit is configured so that when Vbemf ispositive, positive current flows from the power amplifier into the motorwinding. Under these conditions the motor generates positive or motoringtorque. Alternatively, the circuit may be reconfiguration so that whenVbemf is positive, current flows from the motor winding into the poweramplifier and the motor will generate braking torque.

In summary, it will be seen that the invention allows the voltageapplied to the motor windings to be sensed, and from this anapproximation of the back EMF is obtained and this is amplified andprovided to provide the appropriate current to the motor windings. Itwill also be seen that a transresistance amplifier configuration mayalso be employed. Therefore, the winding current may be sensed and usedto provide a signal that has a relationship to the back EMF, and thenthis may be amplified to provide a voltage output to the motor which ismodulated in proportion to the winding current. With the correct choiceof control circuit parameters the winding current can be made to trackthe back EMF with a small phase delay.

It will also be apparent to one skilled in the art that digital forms ofwaveform estimation, determination and prediction may be used accordingto the invention. Thus, from knowing the back EMF waveformcharacteristic for a given motor, it is possible to store waveforminformation in the memory of a microprocessor and from this generatedesired waveforms in real time. Also, a microprocessor may implement thecalculations required to predict the back EMF ie to implement the analogdetermination of the back EMF set forth in example 6 above.

Furthermore, the sensed signals indicative of back EMF may be providedto an amplifier having adjustable gain, which actively controls thecurrent or voltage applied to the motor winding.

ADVANTAGES

The advantages of the present invention have been already mentioned andare principally that changes in speed of a brushless d.c. motor can beeffected with low audible noise and energy wastage. In addition theadvantages include the use of readily available components to achievethis end without resorting to expensive electronic and mechanical meansto improve speed variability and control.

Although the advantages presently benefit brushless d.c. motors of theexternal rotor design, it is envisaged that the same advantages canbenefit motors of other designs by adaptation and modification of thepresent invention by those skilled in the art.

VARIATIONS

Different pole or phase numbers may be used—although this invention isbest suited to small (less than 100 watt) variable speed brushless DCmotors. In the example the motor is a 4-pole 2-phase external rotorradial flux motor—although other configurations may be used. By usingour design of radial flux motor (for example using soft iron wire ortape wound as a thin layer on the stator below the stator windingsdescribed in patent specifications published under our earlier companynames Cadac Holdings Ltd or Cadac Ltd) it is possible to make a lowinductance motor particularly suited to this control technique.

If a sense winding is used a current amplifier may be used in place ofthe voltage amplifier described in the examples.

Finally it will be appreciated that various other alterations andmodifications may be made to the foregoing without departing from thescope of this invention as set forth.

Throughout the description and claims of this specification the word“comprise” and variations of that word, such as “comprises” and“comprising”, are not intended to exclude other additives, components,integers or steps.

What is claimed is:
 1. An electric motor including an electronic backEMF sensing means, the electronic back EMF sensing means including avoltage sensing means and a current sensing means and a rate-of-changeof current sensing means adapted to sense a back EMF generated by arotor of the motor, wherein output signals from the three sensing meansare applied as input signals to a summing amplifier so that an outputsignal of the summing amplifier is analogous to a voltage fieldgenerated by the rotor of the motor and the output signal of the summingamplifier is applied to a power amplifier, which power amplifiergenerates an output which is applied to windings of the motor as a drivevoltage or current, and whereby in use, audible noise associated withthe motor is thus reduced.
 2. An electric motor as claimed in claim 1,wherein the electronic back EMF sensing means also includes filteringcircuits that can introduce a phase shift between the summing amplifierinput voltages and the summing amplifier output voltage which is thenapplied to the power amplifier input.
 3. An electric motor as claimed inclaim 2, wherein the power amplifier comprises power conversion means toadjust an available supply voltage to voltage levels suitable fordriving the motor at the required speed, controlled either directly ofindirectly by the sensing means or the electronic back EMF sensingmeans.
 4. An electric motor as claimed in claim 1, wherein the currentsensing means detects the steady state value of the motor current ineach motor phase.
 5. An electric motor as claimed in claim 1, whereinthe rate-of-change of current sensing means detects the rate-of-changeof motor winding current in each phase.
 6. An electric motor as claimedin claim 1, wherein the voltage-sensing means senses voltage applied tothe motor windings.
 7. An electric motor as claimed in claim 1, whereinthe voltage-sensing means senses voltage applied to the motor windings.8. A method of controlling the signals applied to an electric motor as adrive voltage or current including the steps of: sensing the steadystate current in each motor phase; sensing the rate-of-change of currentin each motor phase; sensing the voltage applied to the motor windings;applying the sensed signals to a summing amplifier so that the output ofthe summing amplifier is analogous to the voltage field generated by therotor of the motor; applying the output of the summing amplifier to apower amplifier; and applying the output of the power amplifier to thewindings of the motor to maintain motor torque and reduce audible noiseassociated with the motor.
 9. An electric motor, comprising: a rotor;and an electronic back EMF sensing means, the electronic back EMFsensing means including a summing amplifier, a power amplifier, avoltage sensing means, a current sensing means, and a rate-of-change ofcurrent sensing means connected to sense a back EMF generated by therotor, output signals from the three sensing means connected as inputsignals to the summing amplifier so that an output signal of the summingamplifier is analogous to a voltage field generated by the rotor of themotor, the output signal of the summing amplifier connected to the poweramplifier, and an output signal of the power amplifier applied towindings of the motor as a drive voltage or current.
 10. An electricmotor as claimed in claim 9, wherein, the electronic back EMF sensingmeans further comprises filtering circuits to introduce a phase shiftbetween input voltages and an output voltage of the summing amplifier.11. An electric motor as claimed in claim 10, wherein the poweramplifier comprises power conversion means to adjust an available supplyvoltage to voltage levels suitable for driving the motor at the requiredspeed.
 12. An electric motor as claimed in claim 9, wherein the currentsensing means detects a steady state value of motor current in eachmotor phase.
 13. An electric motor as claimed in claim 9, wherein therate-of-change of current sensing means detects the rate-of-change ofmotor winding current in each phase.